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Thursday 2 May 2024

Direct Conversion Receive System (Video Supplement)


Above — Drilled chassis for the popcorn direct conversion receiver test bed. AF power amp in place.
The local oscillator and input filter lie in separate boxes and won't be discussed.

Direct Conversion Receive System (Video Supplement)

Greetings. Amidst work, landscaping our property and growing 150 + plants under lights for the gardens, I'm slowly working on a video. The video is Part 2 to The 50 Ω audio preamplifier that goes after a diode ring product detector in DC receivers - Part 1

I've built a lot of stages for this video -- but wanted to test a few of the better 50 Ω input Z audio preamps in an actual DC receiver. A functioning direct conversion receiver provides a good way to test for voltage and power amp instability & noises. I also wanted to get a feel for how much gain we really need despite all the mythos about this topic. With this supplement, I won't have to go into too much detail about the test receiver on the video.

My main receiver goal hasn't changed for 25 years — lift desired RF signals out of the ether with a decent signal to noise ratio while listening to a speaker at comfortable room loudness.

 Above — DC receive system. My chassis contains 2 BNC mixer input ports, an SBL-1 diode ring product detector, a post-product detector network and then 3 audio amplifiers plus a speaker jack. The AF amps = a 50 Ω impedance voltage amp, a second voltage amp with an active gain control - and finally a PA. I'll cover each AF stage separately, but first I'll show the complete receive system from inputs to output. Each stage lies on a separate piece of copper clad board. The active stages are numbered 1, 2 and 3. I soldered 2N4401 or 2N4403 for the BJTs.

Above — My DC (reference) receive system. A clean and simple design. I chose a common base amp for the 50 Ω input impedance first voltage amp  This old & familiar voltage amp seems a great reference piece -- I've built many 10's of them over time. With an added RF band-pass filter, plus LO giving 7 dBm available output power; this seems like a nice piece of kit. All receiver tested 50 Ω input impedance voltage amps in my video will get compared to this reference receive system.
DC and Ugly Construction

Not shown is the DC input. I have a 100 nF cap shunt to ground right on the DC input jack + a 470 µF + a 1 µF capacitor on the main DC line buss *.  The 3 audio boards are floating from the metal chassis and star grounded to a central point on the DC power buss.

* I build with Ugly Construction from DC to ~ 2.2 GHz and employ AF or RF bypass capacitors for DC line standoffs as they feed DC to each stage. E.g. I prefer to not use 1 to 10 megohm resistors as standoffs. Ideal voltage sources are low impedance -- I avoid any resistor unless it's needed & if needed, I strive to keep that resistor value as low as possible for sake of noise hygiene.

No receiver active low pass filtration gets applied as I want to hear all the noise, warts and quonks in my reference receiver. The logical place to insert a low-pass seems the input of the stage marked number 2. The shunt 470 pF cap serves as a simple low-pass filter in my receiver.

My RF bypass capacitors are suited towards lower HF reception. I've got some suggested RF bypass caps from 6.6 to 220 MHz covered here  If you use SMT parts, this chart will be off. I have measured the series resonant frequency of every RF capacitor in my lab -- and have it written in a notebook, or on that part's drawer or envelope.
AF Stages — More Detailed Analysis

Let's follow best practices & show these 3 stages from output to input -- just like I built and tested them.

Above — Schematic of the PA with DC measures. At this point, I did not add the 22 Ω DC decoupling resistor shown in the completed receiver. Few homebrew PA's will oscillate when tested stand-alone with a signal generator into a resistive load. I also took AC measures with my DSO.

A single voltage divider network feeds each of the 2 current sources. A 220 µF capacitor shunts voltage divider noise to ground.

My readers/audience asked me to make a single DC supply audio PA only using TO-92 transistors as finals.This is it. I worked hard to find a solution where the maximum output transfer function would compare with the venerable LM386 -- and bonus -- this final transistor pair tend to not suffer thermal runaway and smoke up your lab.

The key design features to get those goals included serious degenerative feedback [ 68 + 39  + 1 Ω   resistors ], plus current sources to drive the input pair and VAS/final base bias stack. I also set the voltage gain to 21. By increasing the the  2K7 feedback resistor, higher gain lies on tap (a voltage gain of 80 or more arises with a higher feedback resistor); however, this is a power amp and not a voltage amp. Low noise best practices suggest you build up your AF signal voltages with low-noise voltage amps and not within the PA stage.

Above — A close up of the PA in-situ. A temporary orange coloured 1 µF metalized poly film cap lies at ~ 6 o'clock. I connected either a 1 KHz tone or a CD player to the PA via this capacitor. I listened to this PA with my CD player for 4 nights and it sounded lovely & crisp. I built 2 separate PA stages to ensure my design worked. Although preferable, I did not match the input emitter-coupled pair.

Perhaps foolishly -- I did not place heat sinks on my final complimentary pair. All the base drive current comes from the current source and not the usual complimentary pair that drives the finals. Thus they do not run as hot as any other decently designed TO-92 stages I've built.

 Above —  An FFT of the PA driven to 250 mW with a ~ 1 KHz tone.

Above —  An FFT of the PA driven to 697 mW with a ~ 1 KHz tone. Pretty good results from a single 2N4401/2N4403 emitter follower pair.

PA Instability 

Once you connect all your AF stages together in a DC receiver, unwanted audio oscillations may occur.
This might be motor boating — a pulsed, typically low frequency oscillation that may even vary in amplitude and cause squegging. In addition to motor boating -- a steady, higher frequency oscillation tone that sounds hollow "or howls" may arise -- this usually occurs at loud volume.

I learned to think of your DC supply line as a highway connecting various stage inputs to outputs throughout your audio chain. Decoupling the DC line with series resistors and bypassing with AF and sometimes RF capacitors shunt to ground helps to stop AC signals from travelling along this highway. The ultimate way might be to use a capacitive multiplier BJT as shown on the first preamp labelled "one". The capacitor value connected to the base gets multiplied by the Beta of the transistor which sets a long time constant for very low frequency oscillations and those above this low corner frequency.

For motor boating, I normally place a 10-22 Ω decoupling resistor and both a RF and AF bypass capacitor on the PA DC line. I suggest a 470-1000 µF for the DC audio bypass capacitor as a minimum starting value. Each stage in your DC receiver should get some low pass filtration with such an RC network to keep AC signals from travelling down the DC highway.

Further, AF & HF oscillations may also occur in your PA voltage amp called the "VAS". 

AF/RF oscillations also require low-pass filtration, but often just a local bypass capacitor alone will do the job. My PA emitted a ~ 800 Hz howling sound when the volume was turned up loudly. I tamed this by soldering a 270 pF MLCC RF cap from the emitter shunt to ground. For my guitar amps, I've had to apply other strategies.

Above —  Ways to tame audio oscillations in a PA. The emitter degeneration resistor R1 could be increased to lower VAS gain. For example, from 39 to 47 Ω . And/or the C1 value could be increased to get the best result at high drive into the PA. This testing will annoy your family if you listen through a speaker like me! The VAS serves as the main PA voltage amp and offers a big source of instability in some PA designs.


Above — Additional circuitry I've used to tame a 40W guitar amp PA that oscillated at higher drive levels from AF to RF: 100 pF cap from collector to ground. then a 10 Ω resistor on the VAS collector with a RF bypass cap on each base followed by a series R at the BJT base terminal. 

Interestingly in this DC receiver PA build, adding a Zobel network did nothing measurable, so I left it off. I've also connected the VAS base to ground via a series RC network. Sometimes, it's trial and error.

Voltage Amplifier with Active Gain Control

Above — The first version of the inverting active gain control stage. In my reference receiver, I employed the other half of the NE5532 as a follower/buffer. Technically, you do not have to use a buffer, but it helps isolate the active gain stage from the PA input. I've built other active gain control circuits that offer a better log response of the volume control, but this version seems simpler.  If you need more maximum gain, drop R1 to 560 or 470 Ω etc.. 

Active gain controls make sense and may offer headroom and noise advantages over passive volume controls. In a typical Ham receiver, we'll run the 2nd voltage amp at maximum gain and then use a grounded potentiometer either before or after this stage to drop the signal amplitude to a comfortable listening level.

However, the voltage amp always runs at full gain; potentially reducing headroom -- and perhaps more significantly, amplifies the noise by that maximal voltage gain. With an active gain control, noise is amplified only by the exact amount of gain needed to hear a signal.

Above — The sublime FFT of the active gain control with a NE5532. It's very difficult (but not impossible) to get distortion this low with discrete transistors. Further, the device noise performance is much better if you apply a low noise op-amp over discrete BJTs. I'll likely offer a discrete BJT version(s) of this amp in the future. I've designed and built several.

First Voltage Amp — 50 Ω input impedance stage   Number one in the reference receive system
Above — A common base amp biased for 0.71 mA to give a 18.2 dB return loss. I feel it's essential to use an active decoupler/low pass filter - the capacitive multiplier circuit. These are common in industry: ripple filters for the DC supply in VCOs & multiple other products. Roy, W7EL used one in his Optimized QRP Transceiver for QST in August 1980. I built 2 of these back in the day. He standardized using an active decoupler in the first voltage amp for direct conversion receivers. A jolly good thing.

I prefer to follow the common base with a emitter follower: as aforementioned, this proves essential if you plan to use a active gain control with its inverting input. Further, you get maximum gain since the collector is less likely to get loaded down by any stage that follows. With a voltage gain of 107, this stage pretty much sets the noise figure for the receiver.
That's all. Back to the garden.  Best to you!

Sunday 25 February 2024

The 50 Ω audio preamplifier that goes after a diode ring product detector in DC receivers - Part 1

 I posted my longest video to date:

I'll post a few images from the video in the days ahead. Making such a big video felt pretty exhausting.

Best to you

Thursday 4 January 2024

Audio Frequency Return Loss Bridge — 50 Ω —


Above — +/- 15 VDC input and ground ports on die cast chassis.

Above — Side view showing all the input and output ports.

Above — Schematic of 50 Ω differential bridge assembly. I employed a split DC supply to boost headroom and simplify op-amp biasing.I use the moderate power BD139/140 for the filter transistors: a sturdy part with low flicker noise --  no apologies.

Above — Input ports. Left: DC input (direct with a wire) using an SMA connector. Middle: AC coupled port with RCA jack. Built in 220 µF coupling cap allows testing of 50 ohm input Z audio amplifiers with no worries about the bridge causing a DC disturbance of the biasing or current.
Right: 50 Ω audio signal generator input with a BNC connector.


Above — The output of the instrumentation amp U1 gets buffered by the U2a follower. Low impedance output to use a 50 Ω terminated DSO as the detector.


Above — In analog output direct conversion or superhet receivers that use a diode ring product detector, we often employ a simple post product detector network that some refer to as a diplexer. It's not quite a diplexer, although, it does provide a 50 Ω termination to a narrow band of RF frequencies.
You might sweep this network at AF and RF with return loss bridges to study the input match versus frequency.

Above — My current post product detector network with part values chosen to try and match from 200 Hz to 200 MHz. This proved very difficult with such a simple network because the bandwidth is huge and really this calls for 2-3 networks to get it done. However, in simple receivers, this basic network works OK. The impedance match looks terrible from ~ 1 to 4 MHz, however, trying to fix this worsened the match elsewhere.

I performed the above AF measurements with my old audio return loss bridge built in 2010. It failed recently -- and that failure prompted me to design and build this new AF return loss bridge.

Compromise is a key term in simpler RF design. The network components shown gave me the best overall input Z match from 200 Hz to 200 MHz. This network also provided decent low-pass filtration of the RF lurking in the product detector's audio output. A 220 µF (or higher value) audio coupling capacitor helps keep the input noise down in the AF preamp.

Above — A 50 MHz wide sweep of the post product detector network in a tracking generator-spectrum analyzer. The 220 µF capacitor was removed for this RF measure.


Above — Testing gear used in the video: a 50 Ω Mini Circuits SMA terminator + barrel connector to 50 Ω coax -- and an RCA jack with a 2K potentiometer.

 Above — It's always fun to acquire more test gear.

Wednesday 1 November 2023

50.1 MHz VFO + Hybrid Combiner for 2 Tone VHF Testing


Above — This blog post supports the video shown above


Steve AA7U &  Everett N4CY, built gear -- plus a procedure to test Intermodulation Distortion (IMD) on a loop amplifier using a Siglent SDG2042X generator and SSA3021X spectrum analyzer. Click on this hyperlink to read about it.  I'm a fan of Siglent test equipment.

My strategy employs a 50.0 MHz crystal oscillator-based signal generator plus a 50.1 MHz VFO as the second signal source. My VFO tunes from about 49.6 to 51.8 MHz via a front panel air variable capacitor.

My 50.1 MHz VFO

Above — VFO schematic. Although I had worked out the low-pass filter L and C values, I built this VFO without a schematic and perhaps would build it differently if I needed to make another. I might consider tuning the output of the differential amplifier buffer for more output power and less harmonic energy.

I thought mostly about temperature drift when making this -- I started with JFET amp as the oscillator and struggled to make it work. This would be wise since a JFET offers better temperature drift over a BJT and gives a cleaner output signal with lower phase noise.  However, I only had 1 day for this entire project and got frustrated. I deployed a common base PNP BJT local oscillator (LO) that never fails for me.

Both the LO and its buffer get regulated, well filtered DC. The LO gets temperature compensation/separation from the 8.2 volt Zener diode-based voltage regulator by way of 2 R C low-pass filters. I applied several C0G caps to resonate the tank and ran 2 air variable trimmer capacitors -- 1 as the main board frequency trimmer, the other as the front panel tuning control.

The LO gets lightly coupled via 1 pF to a differential amplifier emitter fed 10 mA with a current source. Differential amps offer strong reverse isolation, plus a reduced 2nd harmonic if the BJT balance is OK. The BC546 pair offer reasonable balance right out of the bin (without matching) & the BC546C serves as my go-to differential amp BJT from DC to ~ 100 MHz. The 10 mA current source, plus the 21 mA current in the final feedback amp provide heat for my temperature compensation scheme.

Low-pass filters built using T30-10 toroids worked OK. This was a board cram -- so the inductors are not spaced apart as much as when more board space is available. The 22 gauge air inductor measured ~ 374 nH & seems well anchored to the main 1-sided board with J-B Weld epoxy, plus the grounded coil lead soldered to the main board. The main board =  1/16″ (1.60 mm) Half Ounce 500 Series Copper Clad Board from MG Chemicals.

Above — Copper board under test. To simulate the front panel capacitor, I've got the air variable front panel tuning cap in a small bracket that I got from a local Builders merchant. I have several for holding caps, jacks and potentiometers during test phase circuit development. Mine are all pre-drilled with the proper sized holes to fit pots jacks, or air-variable trimmer caps.

Above — Close up of the tank coil secured with a messy application of epoxy.

Above — Side view. The actual front panel capacitor leads were this long to allow slack to put on the herring tin cover. The Herring Tin lid added much difficulty with temperature compensation and construction tactics -- but I got it done!  The idea of the herring tin cover came from this blog post

Above —View for the VFO showing the DC input port ( an RCA connector ) plus the SMA RF output port. 2 bolts hold the tin to the copper clad board.

Above — My 50.0 MHz xtal based oscillator next to the Herring made VFO. Ready for 2 tone testing. The front panel tuning capacitor is front left. The front panel bolt just fills in a hole I drilled by mistake.

If I want to drive a DUT such as a high IP3 amp -- or say a diode ring mixer ( I rarely use them anymore), I'll chain up 1 of 3 separate, sealed up wide band amplifiers that range from 12 dB to 26 dB gain (up to 150 MHz or so). I also have a plethora of low-pass and band-pass filters in sealed Hammond cases that go from 5 MHz to microwave if needed.


Above — The VHF targeted hybrid combiner is also a return loss bridge and vice versa. No experimenter bench should likely be without a return loss bridge or 3.  I built with standard 1/4 watt 1% metal film resistors and tried several different coils as the transformer. After many versions, I settled with 3 stacked BN61-2402 ferrites with 4 total turns of lightly twisted wire. I twisted the wires only enough so they would stay together during winding. Because of only 4 turns, I was able to use 28 gauge wire. I measured 43 dB port isolation at 50 MHz.

Above —  The applied transformer.

Above — Boxed up combiner/return loss bridge with a Mini-Circuits Lab 50 Ω SMA resistive load attached.

Above —  Another view of the hybrid coupler

Above — My favourite design project of 2014: a wide band return loss bridge with directivity >= 30 dB from 5 MHz to 1.5 GHz.   You may read more about it in the old site archive: Topics 2012 - 2014 : Caitlyn 310 — UHF Beginnings : 3. Return Loss Bridge Experiments : Bridge #4

Tuesday 24 October 2023

Popcorn QRP Audio PA for Receivers and Projects

Greetings!  For 2-3 years, I’ve received emails from readers seeking a simple “popcorn” discrete transistor PA to substitute for the LM386 part in their DYI projects. Readers wanted 3-4 transistors maximum & no differential amplifiers with current sources — and hopefully low distortion up to 1 watt with a ~12 VDC single supply.  

That seemed a tall order, but I did it (more or less). I’ll define ‘popcorn’ to mean that at maximum clean signal power, all harmonics are down to -50 to -55 dBc. This amp behaves well until driven to about 1.3 Watts.  I made a video that lies in the last section.

Above — The final Popcorn QRP PA.  4 transistors. Voltage gain = 28. Quiescent current = 73.5 mA.
This is a power amp designed to cleanly drive a speaker even at loud volumes.  To reduce distortion + boost stability, I applied ample local + global feedback which lowered gain. I suggest readers consider building up their AF signal voltage with low noise, low distortion, feedback-containing voltage amplifiers -- and not rely on their PA stage to make all the voltage + power gain.  Getting most of your voltage gain in your PA adds too much noise into your AF chain.

Above — An FFT of the Popcorn AF PA driven to exactly 1 Watt output power. The load = a 7.9 Ω “resistor” consisting of 3 two watt resistors in parallel. The second harmonic lies ~53 dB down.

Above — LM386 driven to 808 mW. This is the only LM386 scope trace I had where the voltage gain = 40 plus I had applied a good negative feedback network. Therefore, this practice seems a reasonable head-to-head test against the most venerable LM386. The Popcorn PA makes less distortion at 1 Watt, than the LM386 does at 0.81 W.  At 1W power, the LM386 begins compressing into a square wave.

I promote bench experiments – and developed this amp on my bench. I began with a lower power version using 2N4401/2N4403 complimentary emitter followers to drive the speakers. Push- pull drive as opposed to a single-ended PA driver seems the best way to go for decent output power. You might substitute any number of small signal BJTS such as the 2N3904 for the 2N4401 (or the PNP equivalent) in this project.

Let’s start where I began. I’ll show the development of the Popcorn AF PA and give ideas to consider in your own experiments.




Above — The schematic of the initial & fledgling Popcorn PA using paired 2N4401/2N4403 as the complimentary emitter followers.  In 1956 while working for RCA, H.C. Lin developed the first transistor power amp that didn’t use an output transformer. By around 1968, output transformers in solid state AF power amps had all but disappeared in professional designs.

Audio transformers suffer from non-linearity and in the case of the tiny transformers employed in cheap transistor radios of lore — these gave distortion, poor bass response -- plus very low output unless run in push-pull fashion. I suggest there are < 2 coherent reasons to use AF output transformers for solid state designs in 2023.
Input Stage

Without a differential pair as the input stage, I chose a PNP for the Q1 input amp with global negative feedback coming from the output rail going back to the Q1 emitter. The Q1 emitter also gets local feedback -- AC degeneration through the 330 Ω resistor. Because of all the feedback on Q1, Q2 provides most of the voltage gain and gets around double the collector current.

In all PA versions, Q1 bias gets set by a potentiometer (20K here). The pot proves necessary since all of us use a slightly different DC power supply voltages. The potentiometer allows you to optimize the Q1 bias for the lowest possible distortion with whatever DC power supply you use. When satisfied, you may remove the pot, measure it, and replace it with 1 -- or 2 series or parallel resistors to try to get as close as possible to the measured pot value. Alternately, you hard wire in a 20 – 25K trimmer potentiometer.  
In the final Popcorn PA version, I show a fixed Q1 bias resistor and a procedure how to set this value

The Q2 “stack” includes Q2 & all the parts connected to the Q2 collector going straight up to the positive DC power supply rail. Q2 serves as the main voltage amplifier. I placed a 10 Ω emitter resistor as local negative feedback to stabilize the stack against HF during development. I have not found any HF instability in the Popcorn PA with or without that 10 Ω resistor.

With the 2K Q2 collector resistor, the stack draws ~ 2.5 mA. Let’s look at some DSO outputs:

Above — DSO time domain output. The first draft PA driven to 2.01 volts peak-peak. Lovely sine wave.  Power = 64 mW.

Above — The FFT of the PA driven to 2.01 Vpp or 64 mW into a 7.9 Ω load.

Above — Left PA driven to 4 Vpk-pk [ 253 mW ] and 5 Vpk-pk [ 396 mW ]. Only the fundamental 2nd,3rd and 4th harmonics shown.  The 3rd harmonic tone starts to rise as the amp is driven to 4 Vpp. You can see the limitations of a single pair of TO-92 transistors such as the 2N4401/2N4403.

We’ve already exceeded the harmonic distortion goal for a popcorn PA amplifier. That is --- all harmonics must be down 50-55 dB at the maximum clean power

Above — FFT with PA driven to 6 Vpk-pk or 570 mW.  The 3rd harmonic is only 27-28 dB down. These TO-92 transistors are getting hot and starting to stink. Some of this distortion might be Beta droop from the high collector current plus heat.

Regardless, this seems like unacceptable distortion. You could easily hit power level this high on a strong Morse code (CW) station.
At this point, the 2N4401/4403 emitter followers seem only good enough for headphone level listening.

What can we do to try boost their linearity?

Technique One — Bootstrapping

Above — Boot strapping Q2.

Q2’s 2K collector resistor gets split to make a tap for a 330 µF bootstrap capacitor that provides positive phase feedback from the output rail to the collector. This raises collector impedance and reduces the loading effects of the Q2 collector resistance on the input of our 2 complimentary emitter followers. The positive feedback lowers Q2 signal drop.

Above — The FFT of the PA driven to 2.0 Vpp or 63 mW into a 7.9 Ω load. If anything, the 3rd harmonic is about the same while the rest are a bit worse. Bootstrapping is not helping here.

Above — The FFT of the PA driven to 4 Vpk-pk or 253 mW . The third harmonic is about the same without bootstrapping, while the other tones look a bit worse.

Above — FFT of the PA driven to 6.03 Vpp or 753 mW.  In this case, the harmonic distortion has improved. For example the 3rd harmonic improved by about 7 dB.  But overall, the net distortion exceeds our harmonic distortion goal.

Theoretically, bootstrapping may help and often works as well as driving the Q2 stack with a current source.  However, it doesn’t seem to work in this simple amplifier with a 2N4401/2N4403 pair.

Above — A fun FFT of what happens when you submit the 2N4401/2N4403 pair to 1 Watt power. Lots of compression, square waves & those emitter followers are smoking hot + stinking up the room.

Technique Two — Current Source

Above — I biased a single PNP to function as a current source. I set the output current as close as possible to that of the Q2 stack with the 2K collector resistor (limited by standard value resistors). The current source provides high impedance drive to the emitter follower pair. I won’t show any tracings because the current source, like the positive feedback, didn’t reduce distortion --- and in for some tones, worsened it.   I went back using a collector resistor.

Technique Three — Reducing the 2K collector resistor to 1K Ω

With the 2K collector resistor, the stack current measured ~ 2.5 mA. I measured the Q2 stack current at 4.83 mA when reducing the 2K Ω resistor in half. The results seemed unimpressive.

Above — For reference, With the 2K collector resistor driven to 3 Vpp.  [142 mW power]

Above — With 1K Q2 collector resistor driven to 3 Vpp. The 2nd harmonic improved by ~ 5 dB and the 3rd by about 4 dB.   At higher power like 500-600 mW, , the distortion was still too high for my liking. Further, the increase in amplifier quiescent current for the net reduction in harmonic content wasn’t worth it.

I’ve gone as far as I can with the simple 2N4401/2N4403 emitter followers. I’ve got to add some current gain and get some proper power followers. 

Before, we go to Section 3, the high power version of the Popcorn QRP PA -- Section 2 quickly covers output stage biasing:

----   [ SECTION 2  ]  OUTPUT STAGE BIAS   ----

2 diodes produce a voltage drop of around 1.3 volts providing sufficient bias for the 2N4401/2N4403 output emitter followers. From reading & my own experiments, the output bias may affect PA output distortion. The most obvious way is by giving crossover distortion.

Above — DSO screen capture of the low-power Popcorn PA with only 1 bias diode across the emitter follower bases. You may easily see (and hear) crossover distortion.

Above — An FFT of the 1 diode output bias with only the amp driven to 36 mW output power. The distortion dominates with odd order harmonics.

Above — FFT after adding back the 2nd output bias diode. This reduced the amplifier distortion shown above. Crossover plus output follower switching distortion pose factors we must live with. How far the output pair are biased from Class B towards Class A may also affect amplifier distortion.

However, using 2 diodes, we don’t have much control over that. You may place a small value resistor in series with 1 diode instead of using 2 diodes -- or in series with 2 (or more) diodes to change the output bias. An alternate way is to remove the diodes and replace them with a transistor.

Above — Schematic with the 2 diode bias replaced with an NPN referred to as an amplified diode or Vbe multiplier bias generator. Normally, this BJT has a trimmer resistor as R1 in the schematic for tweaking the voltage divider bias. The trimmer gets adjusted while watching the output in a test circuit to find the sweet spot of bias -– the setting that offers the lowest distortion in the output. 

I normally temporarily make R1 a trimmer pot, set the bias and then remove and measure the trimmer pot. Then I replace that with a fixed resistor such as the 1K8 Ω shown.

Since this is the popcorn PA stage, we’ll stick to plain diode biasing of the output followers.


Above — Device under test. The best part about bench building is getting to use your test equipment. Glory and fun on the bench. Since I usually make 22 – 50 watt PAs, my electrolytic capacitor collection are all rated 50 volt to 100 volts. They look quite large in the Popcorn PA.

Above — Popcorn PA with DC voltages. Q1 shows fixed bias. I’ll give the bias procedure soon. The 10 Ω Q2 emitter resistor got dropped since this adds 1-2 dB of lower tone harmonic distortion under heavier drive.

I didn’t bother with the standard Zobel network in parallel with the speaker as seen in most AF PAs. This series cap + 10 Ω resistor across the speaker serves to lower the Q of the resonant peak of the speaker’s peak impedance at somewhere between 80 and 130 Hz.  While important for crossover design + function, I’ve left it out. You may need it with some speakers perhaps.

Power Followers

Above — I swapped out the TO-92 finals for some big boots.  In many lower power amps, to get current gain you’ll keep the TO-92 followers and then drive another set of power followers such as the BD139/BD140 pair. This works well and is recommended, however; we’re going full on popcorn on this project.
Thus, we’re keeping the emitter follower driving an emitter follower theme, but combining both in a packaged Darlington pair. This keep the parts count down -- plus provide the high Beta and current we seek to drive our speaker with room filling, low distortion loudness.

The TIP122/127 pair are only 1 example of packaged Darlington current amplifiers. I’ve got 3-4 other in my parts bins such as the BD94C/93C or TIP142/147 pairs – but again, usually I build higher power amps.

I bought the TIP122/127 pair for $2.30 Canadian dollars & they look husky and tough. You don’t even need to heat sink them for 12 VDC power.  If you need to heat sink them, then it's easy to do. Some readers emailed me to say they had smoked countless 2N3904/2N3906 pairs in their PA building adventures. Some soldered several in parallel to make a "power follower", etc..  

While purists may dislike a packaged Darlington pair – they seem perfect for popcorn PA stages and practice guitar amps alike. We have to add 2 more diodes to properly bias both Darlington transistors.

I added the Q2 boot strap back in. For this version, it significantly helped boost linearity from low to high power.

I kept the 1 Ω emitter resistors of the low power version. In pro audio, these are usually 0.1 or 0.22 Ω but of course, those amps are making some serious power.
In the past, I placed two or three quarter watt 1 Ω resistors of 1% tolerance in parallel to get the maximum possible output power. I left the popcorn emitter resistors at 1 Ω to ensure this project is stable for anyone choosing to experiment with it.

Play with every resistor value on the test bench. You’ll probably make a better PA than I did.

Let’s go through some FFT’s of the Popcorn PA at various drive levels:

Above — FFT at 3 Vpp. This proved the lowest 2nd harmonic tone measured @ -56 dBc. You could further experiment with the output bias, add a current source, or perhaps make other tweaks, if chasing a lower 2nd harmonic tone is your goal.

Above — FFT with the Popcorn PA driven to 6 Vpp or 570 mW.  Looks about the same as with 3 Vpp.

Above — FFT at 7.5 Vpp. Again it look similar to the Vpp = 3 or 6 FFTs.

Above — Cranking up the drive! FFT while driven to output 8.39 or 1.11 Watts. Still meets our popcorn goal of all tones down 50-55 dB at maximum clean power.

Above — FFT while driving the PA to 9.18 Vpp.  The harmonic tones are starting to rise!

Above — FFT while driven to 1.34 Watts. Things are falling apart.  Ok, let’s finish up.

Above — Set up schematic.  If your power supply is close to 12 VDC, then consider just building the fixed Q1 bias version shown earlier. However, bigger is better in PA stages. If you’ve got 13.8 or 14 VDC, then your maximum clean output power will go up. You may choose to optimize Q1 bias for a different DC supply.

In big power PA’s the DC rails are often unregulated. Fortunately, most of our ~ 12 VDC single power supplies are regulated which makes setting up the Popcorn AF PA a snap.

Terminate the output with a 10 ohm or lowish value resistor – or your bench 8 Ω load. Do not connect anything to the input.  Preset to maximum resistance, connect a 10 - 25K pot from the DC power supply rail to the Q1 base. Clip your positive voltmeter probe to the output rail and tweak the pot until your measured DC = your DC supply voltage divided by 1.82. Remove the pot and measure. Substitute the nearest standard value or place 2 in series or parallel to get close to this voltage.

If the output rail voltage lies between VCC/1.82 & VCC/2 you’ll be fine. Of course, you may experiment to find the optimal Q1 bias for your particular build -- that serves as the best way to optimize linearity.


I made a short video so you can hear the Popcorn PA in action. I connected it to a CD player plus my 8 inch guitar speaker and cranked it loudly to show its linearity under heavy audio drive.

I sampled at 44.1 KHz into mono using a Lewitt LCT 440 large-diaphragm condenser mic --- the same mic I use for my voice overs. I like the LCT 440 since it offers a flat bandwidth + very low added noise at a reasonably low cost.

Above — It seems better to watch this video on YouTube directly.


To clarify, I think the LM386 is an awfully good part. Imagine if your design team made a linear IC that went into hundreds of thousands of projects or products?  I'm a fan of the LM386 and the designers left us IC pins to add negative feedback with.

I cover this in the following blog post:

Link to my LM386 Experiments from November 2022